Biasing scheme for large format CMOS active pixel sensors

ABSTRACT

An image sensor includes circuitry compensating for voltage drops in a V SS  line. The image sensor includes a plurality of photoreceptors arranged in a pixel array having a number of column lines, and read-out circuitry on the column lines. The read-out circuitry provides substantially equal currents on each column line so as to compensate for voltage drops in the V SS  line and provide more accurate pixel signals. The image sensor also includes circuitry for filtering noise from a voltage supply line, and for providing hard and/or soft reset operations.

This is a divisional of U.S. patent Ser. No. 10/409,108, filed on Apr. 9, 2003 now U.S. Pat. No. 7,408,577, the disclosure of which is incorporated by reference in its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates to active pixel sensor arrays and, more particularly, to a biasing arrangement for complementary metal oxide semiconductor (CMOS) active pixel sensor arrays.

2. Description of the Related Art

Active pixel sensor (APS) imaging devices are described in U.S. Pat. No. 5,471,515, for example. APS imaging devices include an array of pixel cells, generally arranged in rows and columns. Each pixel cell includes a photodetector (e.g., photodiode) that converts light energy into electric signals. The pixel cell also includes one or more transistors. The transistors typically provide amplification, read-out control, and reset of the photodetector signal, and operate to provide reset signal and a photodetector pixel signal as output signals from the pixel cell.

The architecture of a conventional imaging device with a CMOS APS 2 is illustrated in FIGS. 1 and 2. APS 2 includes a row and column array of pixel cells 4. As illustrated in FIG. 1, array rows are oriented horizontally, and array columns are oriented vertically. The 4×4 array pictured in FIG. 1 is merely illustrative. Typical APS arrays are fabricated as much larger arrays.

Referring more specifically to FIG. 2, a representative three transistor (3T) pixel cell 4 is shown as including a photodiode 6 (PD) and a readout portion which includes a source follower transistor 8 (M_(D)) and a row select transistor 10 (M_(SEL)). Pixel cell 4 has a drain of source follower transistor 8 (M_(D)) connected to voltage line 12 (V_(AA) _(—) _(PIX) _(—) _(COL,j)). The source of the row select transistor 10 (M_(SEL)) is connected to a column line 14 of the cell array to which a load transistor 16 (M_(LD)) also is connected. The load transistor 16 (M_(LD)) acts as a current sink for a bias current I_(BIAS) _(—) _(PIX). Thus, each pixel 4 includes a source follower arrangement of transistors 8, 10, with the pixel and a common current sink transistor 16 for each array column.

The operation of APS 2 will be described with reference to the timing diagram of FIG. 3. The photodiode 6 is reset by pulsing a reset transistor 18 (M_(RS)) with a reset pulse Φ_(RS). During the ensuing integration (exposure), the photodiode 6 voltage is decreased by the photo current. After the pre-determined integration period, a row select pulse Φ_(SEL) turns ON and an output signal appears at the node denoted by V_(OUT) (FIG. 2).

The output signal voltage is given by Equation 1, as follows: V _(OUT,sig) =A _(V)·(V_(RS,pix) −ΔV _(sig)−(V _(th) −δV _(th)))  (1) where A_(V), V_(th), V_(RS,pix) and ΔV_(sig) are the voltage gain (A_(V)) and the threshold voltage (V_(th)) of the source follower, the initial photodiode voltage (V_(RS,pix)) just after the reset, and the voltage swing (drop) (ΔV_(sig)) caused by the accumulation of the signal electrons on the photodiode, respectively.

V_(th) represents a deviation of the effective threshold voltage, which gives rise to fixed pattern noise (FPN) in the array. FPN (also called nonuniformity) is spatial variation in pixel output values under uniform illumination due to device and interconnect parameter variations (mismatches) across the sensor. It is fixed for a given sensor, but varies from sensor to sensor. FPN increases with illumination, but causes more degradation in image quality at low illumination. CMOS (APS) sensors have higher FPN than charge-coupled devices (CCDs) and suffer from column FPN, which may appear as shadow stripes in the image and can result in image quality degradation.

The output signal voltage V_(OUT,sig) for a column pixel is sampled and held on a first capacitor in a sample and hold (S/H) circuit 20 (FIG. 1). Then the photodiode is reset again, and the output reset voltage is sampled and held on a second capacitor in sample and hold (S/H) circuit 20.

The output reset voltage is given by Equation 2, as follows: V _(OUT,rs) =A _(V)·(V_(RS,pix)−(V _(th) −δV _(th)  (2) The signal voltage swing, representing the light-induced signal on the photodiode, can be extracted by subtracting (1) from (2), which yields: V _(SIG) =A _(V)·ΔV_(sig)  (3) Thus, variations in V_(RS,pix) and V_(th) are eliminated, in principle.

Referring again to FIG. 1, a row select circuit 22 selects a row to be reset or readout. All rows are selected for read out in sequential fashion. The S/H circuit 20 performs sample and hold operations for the reset and pixel signals for each of the array columns, which are represented by equations (1) and (2). The outputs of the S/H block 20 are fed to a differential amplifier which subtracts the two signals to produce V_(SIG) in accordance with equation (3). This signal is then digitalized and sent to an image processor with other signals from the pixels of array 2.

The FIG. 3 pulse timing diagram illustrates the operation of the image sensor shown in FIG. 1 having photodiode active pixels. Conventionally, the row select and sample-and-hold operations are performed in a row-by-row fashion, represented in FIG. 3 by these successive rows i−1, i, and i+1. A period at the start of each frame between the end of the last row select and the beginning of the first row select is called the vertical blanking period (V_BL).

In a large format array based on the representative architecture shown in FIG. 1, a problem occurs due to the voltage (I-R) drop along the V_(SS) line, which couples ground to all the column circuit sink transistors 16 as illustrated in FIG. 4. During the readout period, voltage drops along the V_(SS) line, which is typically grounded at terminal AV_(ss), cannot maintain the established (ground) level for each column line due to parasitic resistances R_(SS) denoted in FIG. 4. The graph of FIG. 5 illustrates the voltages on column lines 1 to j_(max) as shown, those column lines farthest from terminal AV_(ss) have a voltage above ground while those closest to AV_(ss) have voltages at or near ground along the line V_(SS).

The voltage drops along the line V_(ss) impact the bias currents I_(BIAS) _(—) _(PIX) produced by column line transistors 16. The bias current is given by Equation (4), as follows: I _(BIAS) _(—) _(PIX,j)=β/2·(V _(GS,j) −V _(TH))²=β/2·(V _(LN) −V _(SS) _(—) _(PIX) _(—) _(COL,j) −V _(TH))²  (4) where β, V_(GS,i), and V_(SS) _(—) _(PIX) _(—) _(COL,j) are a process and size dependent parameter (β), the gate-source voltage for the j-th load transistor (V_(GS,i)), and the V_(SS) voltage at the column J (V_(SS) _(—) _(PIX) _(—) _(COL,j)), respectively. Thus, as V_(SS) _(—) _(PIX) _(—) _(COL,j) becomes higher than V_(SS), the bias current decreases. The change in the output voltage due to the higher V_(SS) is given approximately by Equation (5) as follows:

$\begin{matrix} {{\Delta\; V_{OUT}} = {{\sqrt{\frac{\beta_{LD}}{\beta_{D}}} \cdot \Delta}\; V_{SS}}} & (5) \end{matrix}$ where β_(LD), β_(PD), V_(SS) are β for M_(LD) and M_(D), and the voltage change on the V_(SS) line, respectively. If a fixed pattern noise (FPN) suppression operation, where the signal voltage swing is extracted by subtracting Eq. 1 from Eq. 2, is not performed, shading appears in a reproduced image. By applying such an FPN suppression operation, shading caused by the voltage change on V_(SS) line, expressed by Eq. 5, may be suppressed. However, other shading will still be present.

Shading may be caused, for example, by decreased bias currents due to the reduced effective gate-source voltage V_(GS) of the load transistor 16 (M_(LD)), which may result in variations in time constants for charging and discharging the sample and hold capacitors C_(SHS) or C_(SHR) within the sample and hold circuit 20. The capacitor C_(SHS) and C_(SHR) respectively sample and hold the pixel output signal and reset output signal. The time constants are given approximately by Equation 6, as follows:

$\begin{matrix} \begin{matrix} {\tau = \frac{C_{SHX}}{g_{m}}} \\ {= \frac{C_{SHX}}{\sqrt{2 \cdot \beta \cdot I_{BIAS\_ PIX}}}} \end{matrix} & (6) \end{matrix}$ Thus, the resulting voltages on the sample and hold capacitors may change when the pulse width of the sample-and-hold pulse is comparable to the time constant. The pulse width of the sample-and-hold pulse may be comparable to the time constant since higher resolution image sensors require shorter pulse widths to obtain a given frame rate. Another related concern is the source follower gain variation due to the decreased bias current.

Similar voltage drops occur along a voltage line 24 which supplies the pixel voltage V_(AA) _(—) _(PIX), as a result of parasitic resistances R_(AA) along that line, as shown in FIG. 4. The voltage changes here affect the pixel output voltage little, however, since the driver transistor 8 (M_(D)) of each pixel cell (FIG. 2) operates in the saturation region and thus the drain voltage does not affect its drain current. A problem arises, however, during the reset operation.

There are two modes of pixel reset operation, “hard” and “soft.” A “hard reset” refers to a reset where the reset transistor 18 (M_(RS)) operates in its linear region. The initial voltage of the photodiode V_(RS,pix) is given by Equation 7, as follows: V _(RS,pix) =V _(AA) _(—) _(PIX) _(—) _(COL)  (7)

To make reset transistor 18 (M_(RS)) operate in its linear region, a pulse height of the reset pulse Φ_(RS) should be higher than V_(AA) _(—) _(PIX) _(—) _(COL) by V_(th), where V_(th) is the effective threshold voltage of the reset transistor 18. Noise associated with the ‘hard’ reset is given approximately by Equation 8, as follows: v _(hard) ² =kT/C _(PD)  (8) where k is the Boltzmann's constant, T the absolute temperature in K and C_(PD) is the capacitance of photodiode 6, respectively.

A “soft reset” refers to a reset where the reset transistor 18 (M_(RS)) operates first in its saturation region, and then in the sub-threshold region. An initial voltage V_(RS,pix) of photodiode 6 is given by Equation 9, assuming the pulse height of the reset pulse Φ_(RS) is equal to V_(AA) _(—) _(PIX,) as follows: V _(RS,pix) =V _(AA) _(—) _(PIX) _(—) −V _(th)  (9)

When the pulse height of the reset pulse Φ_(RS) is equal to V_(AA) _(—) _(PIX), reset transistor 18 (M_(RS)) operates in its saturation region. The final stage at the reset operation is dominated by the sub-threshold operation. Reset noise associated with the ‘soft’ reset is given approximately by Equation 10, as follows: v _(soft) ² =kT/2C _(PD)  (10)

Although the noise associated with the soft reset is less than that associated with the hard reset, there are other concerns with the soft reset. For example, the soft reset yields non-linearity in low light level photo-conversion characteristics of the pixel cell. The soft reset also yields image lag due to the fact that the final stage at the reset operation is dominated by the sub-threshold operation of the reset transistor 18. Therefore, the hard reset is preferable to the soft reset, especially for motion picture applications, for example.

In order to perform the hard reset, the pulse height of the reset pulse Φ_(RS) should be higher than V_(AA) _(—) _(PIX) _(—) _(COL) by V_(th), which requires an additional voltage pump circuit in a row select pulse generation circuit (assuming that V_(AA) _(—) _(PIX) is the highest voltage supplied to the image sensor.) There may be a case, however, in which the soft reset is applicable when noise performance and dynamic range are the most important performance parameters.

If a large format array is built based on the architecture shown in FIG. 1, a problem associated with the hard reset occurs due to the voltage (I·R) drops along the V_(AA) _(—) _(PIX) line. When a row is selected for readout, the pixel bias currents I_(bias-pix) flow from V_(AA) _(—) _(PIX) to AVSS as shown in FIG. 4. During this row select period, voltages along the V_(AA) _(—) _(PIX) line cannot maintain an appropriate level, as shown in FIG. 6, due to parasitic resistances R_(AA) along the V_(AA) _(—) _(PIX) line.

The pixel reset Φ_(RS) is applied after the signal voltage sampling (Φ_(SHS)=‘H’) while the row select pulse Φ_(SEL) is kept on, as shown in the timing diagram of FIG. 3. Since the pixel reset voltage, which is a drain voltage of M_(RS), is V_(AA) _(—) _(PIX), as expressed by Eq. 7, the initialized voltage of each photodiode V_(RS,pix) on a row is not identical, as shown in FIG. 6. In addition, if the V_(AA) _(—) _(PIX) line is contaminated by noise, that noise is sampled on a photodiode.

BRIEF SUMMARY OF THE INVENTION

The present invention provides readout circuitry for each column of an APS device array in which substantially equal sink currents exist on each column line or in which sink currents are not employed.

In a first exemplary embodiment of the invention, a capacitor loaded source follower transistor is provided for each pixel so that no bias currents are needed. In a second exemplary embodiment of the invention, a level shift circuit is associated with each column line load transistor to apply a constant gate-source voltage to the load transistor. In a third exemplary embodiment a clamp circuit is associated with each column line. In a fourth exemplary embodiment, a differential amplifier is associated with each column line to sink a constant current.

In another exemplary embodiment, a transistor is placed between the line V_(AA) _(—) _(PIX) and the column line V_(AA) _(—) _(PIX) _(—) _(COL). By making the transistor, preferably an n-MOS transistor, operate in its saturation region, noise on the V_(AA) _(—) _(PIX) line can be filtered out. Further, by adding a p-MOS transistor in parallel with the n-MOS transistor, a more reliable hard or soft reset operation of the pixel cell can be obtained.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other features of the invention will become more apparent from the detailed description of exemplary embodiments of the invention given below with reference to the accompanying drawings, in which:

FIG. 1 is a block diagram of a conventional CMOS APS;

FIG. 2 is a schematic illustrating a pixel configuration of the conventional CMOS APS of FIG. 1;

FIG. 3 is a pulse diagram for the CMOS APS of FIG. 1;

FIG. 4 illustrates current paths and parasitic resistances in the conventional CMOS APS of FIG. 1;

FIG. 5 is a graph illustrating voltage drops along the V_(SS) line in the conventional CMOS APS of FIG. 1;

FIG. 6 is a graph illustrating voltage drops along the line V_(AA) _(—) _(PIX) in the conventional CMOS APS of FIG. 1;

FIG. 7 is a circuit diagram of a pixel array having capacitor loaded source follower read-out circuitry according to a first exemplary embodiment of the invention;

FIG. 8 is a pulse timing diagram for the embodiment of FIG. 7;

FIG. 9 is a circuit diagram illustrating a modification to the embodiment of FIG. 7;

FIG. 10 is a pulse timing diagram for the embodiment of FIG. 9;

FIG. 11 is a circuit diagram of a read-out circuit for a pixel array according to a second exemplary embodiment of the invention;

FIG. 12 is a circuit diagram of a read-out circuit for a pixel array according to a third exemplary embodiment of the invention;

FIG. 13 is a pulse timing diagram for the embodiment of FIG. 12;

FIG. 14 is circuit diagram of a read-out circuit for a pixel array according to a fourth exemplary embodiment of the invention.

FIG. 15 is a circuit diagram of a reset circuit for filtering noise on a pixel cell according to an aspect of the invention.

FIG. 16 is a circuit diagram of a selective hard or soft reset circuit on a pixel cell according to an aspect of the invention; and

FIG. 17 is a schematic block diagram of a system that includes an image sensor chip with compensating circuitry according to another exemplary embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

Referring initially to FIG. 7, an APS 30 according to an exemplary embodiment of the present invention is shown. APS 30 includes a number of pixels 34 arranged in an array 32. Pixels 34 may employ many different architectures, including the 3 transistor (3T) architecture shown in FIG. 2 or 4 transistor (4T) or higher order transistor architectures known in the art. APS 30 does not use the FIG. 2 current sink transistor 16, but rather employs a capacitor loaded source follower formed by connecting the column lines 44, 46, 48 of the array to sample and hold circuit 36 in the manner shown in FIG. 7. Consequently, the load transistor 16 of FIG. 2 is eliminated from the circuit. For clarity, certain features of APS 30 not critical to the invention have been eliminated in the drawing of FIG. 7.

Column read-out circuitry of this embodiment will be described for column 48 as being representative of other columns in the array. Connected to column line 48 is a read-out circuit 51 provided in S/H circuit 36. The read-out circuit 51 includes two capacitors 52 (C_(SHS)) and 54 (C_(SHR)) connected in series with column line 48. These two capacitors effectively replace the load transistor 16 (M_(LD)) shown above in the conventional arrangement of FIG. 2, forming two capacitor-loaded source follower circuits with transistor 8 (M_(D)), connected to capacitor 52 (C_(SHS)) or 54 (C_(SHR)) through column select transistor 10 (M_(SEL)).

Signal acquisition and reset of the read-out circuit are controlled by a sample and hold signal Φ_(SHS), supplied to switch 56, a sample and hold reset signal Φ_(SHR), supplied to switch 58, and a capacitor reset pulse Φ_(RSC) supplied to switches 60 and 62.

Operation of the embodiment of FIG. 7 is described below. Although the description is for the conventional PD APS pixel cell 4, shown in FIG. 2, any other pixel cell structure can be used.

Just before the sample and hold operation for the pixels on any one of the rows i, the hold capacitors 52 (C_(SHS)) and 54 (C_(SHR)) are reset to ground (V_(SS)) by turning Φ_(RSC) on. Then the row select pulse Φ_(SEL,i) is turned on (see FIG. 8 timing diagram). When the sample pulse for signal Φ_(SHS) is turned on, switch 56 closes and the pixel output V_(OUT,sig) is sampled onto capacitor 52 (C_(SHS)). After Φ_(SHS) is turned off, the pixels on the i-th row are reset by the reset pulse Φ_(RS,i). Then, the sample pulse for reset signal Φ_(SHR) is turned on and switch 58 closes, and the reset signal V_(OUT,rs) is sampled on capacitor 54 (C_(SHR)). Lines 53 and 55 represent output lines to other circuitry which processes the signals held on capacitors 52 and 54.

The signal voltage V_(OUT,sig) and V_(OUT,rs) are given by Equations (1) and (2), above. In this capacitive loaded source follower, the hold capacitors 52, 54 are charged up to a voltage level expressed by Equations (1) or (2). With this arrangement, the current flow through the driver transistor 8 (M_(D)) of a pixel does not cause current flow and thus voltage drops on the V_(SS) line, which affect the sampled and held signals.

FIG. 9 shows a modification of the embodiment of FIG. 7 in which a line 64 connects column line 48 to ground (V_(SS)) by way of a switch 66 controlled by a signal Φ_(RS) _(—) _(COL). A timing diagram is shown in FIG. 10. The arrangement of FIG. 9 allows the use of a reset signal Φ_(RS) _(—) _(COL) to reset column line 48 to ground (V_(SS)), before the sample-and-hold pulses are turned on and the row select pulse is applied.

Referring to FIG. 11, another exemplary embodiment of the present invention is shown in which a level shift circuit 52 is used to maintain a constant gate-source voltage as applied to a load transistor 50 (M_(LD)). FIG. 11 shows a block diagram of a column readout bias portion of an APS, such as the conventional APS described above in connection with FIGS. 1 and 2. Load transistors 50 (M_(LD)), corresponding to load transistor 16 in FIG. 2, are arranged with a level shift circuit 52 inserted between the V_(SS) line and the gate of each load transistor 50 (M_(LD)). The bulk (p-well) of each load transistor is connected to its source.

The input of the level shift circuit is V_(SS) _(—) _(COL) _(—) _(i) and the output of is V_(SS) _(—) _(COL) _(—) _(i)+ΔV, where ΔV is a pre-determined voltage. With this circuit configuration, the gate-source voltage and the gate-bulk voltage are kept constant even if the V_(SS) line voltage raises due to the I-R drop mentioned above. Thus, the bias current for source follower circuits (which are formed by transistor 8 (M_(D)) and load transistor 50 (M_(LD)) through transistor 10 (M_(SEL))) can be kept constant.

Referring to FIGS. 12 and 13, another exemplary embodiment of the present invention is illustrated in which a clamp circuit 63 is employed. The circuit diagram of FIG. 12 shows a column readout bias portion of an APS, and is similar to the embodiment described above in connection with FIG. 11, the level shift circuit 52 having been supplanted by a clamping circuit 63. FIG. 13 is a pulse timing diagram for the circuit of FIG. 12.

Each clamping circuit 63 includes a clamping capacitor 64 (C_(CL)) connected between the gate of a load transistor 66 (M_(LD)) and the V_(SS) line. The gate side of clamping capacitor 64 also is connected to a voltage line V_(CL) by way of a switch 68 controlled by a signal Φ_(CL).

Referring to the timing diagram of FIG. 13, during the vertical blanking period when no currents flow from V_(AA) _(—) _(pix) to V_(SS) (FIG. 2), the clamp pulse Φ_(CL) is turned on, thereby charging the clamp capacitor 64 at the clamp voltage V_(CL). This configuration makes the gate-source voltage and the gate-bulk voltage constant even if the V_(SS) voltage rises due to the I-R drop when a row select pulse turns on.

Although the clamp pulse Φ_(CL) is applied during the vertical blanking period in FIG. 13, it is apparent that the clamp pulse can be applied any other time when the row select pulses are turned off.

Referring to FIG. 14, a simplified circuit diagram of another exemplary embodiment of the invention is shown in which a differential amplifier 70 is placed on each column readout line. The inputs for the differential amplifier 70 are bias voltages, V_(REF1) and V_(REF2), and the output of each differential amplifier 70 is connected to the column line. The differential amplifier is powered by V_(AA), which preferably is different from V_(AA) _(—) _(pix).

The differential amplifier is designed so that it draws current, which is expressed by Equation 11 as follows: I=I ₂(V _(REF2))−I ₁(V _(REF1))  (11) and voltage change or noise on V_(SS) line does not affect the value of the current I.

The proposed embodiments above are especially useful for large format image sensors since identical bias currents for columns can be obtained. Shading in a reproduced image, which is caused by I-R drop on V_(ss) line, can be eliminated.

Further modifications to the above-described exemplary embodiments of the invention provide for filtering of a pixel reset signal and provide for the ability to operate a ‘hard’ or ‘soft’ reset. An embodiment is shown in FIG. 15, in which an n-MOS transistor 80 (M₁) is connected between the V_(AA) _(—) _(pix) line and the drain of M_(D) and M_(RS) of a pixel. The gate of transistor 80 (M₁) connects to a bias line V_(AA) _(—) _(PIX) _(—) _(CTL) which is usually biased at the same value as V_(AA) _(—) _(PIX) so that transistor 80 (M₁) operates in its saturation region. Since no currents flow on the V_(AA) _(—) _(PIX) _(—) _(CTL) line, the gate voltage of transistor 80 (M₁) is maintained constant. On the other hand, voltages along the V_(AA) _(—) _(pix) line change due to the IR drop during the period when Φ_(SEL) is turned on as mentioned above in connection with FIG. 6.

The source voltage of transistor 80 (M₁), which is V_(AA) _(—) _(PIX) _(—) _(COL) _(—) _(j) is described by Equation 12, as follows: V _(AA) _(—) _(PIX) _(—) _(COL) _(—) _(j) =V _(AA) _(—) _(PIX) _(—) _(CTL) −V _(th)  (12) where V_(th) is the threshold voltage of transistor 80 (M₁). Thus, the source voltage is constant regardless of the voltage variation or possible noise on the V_(AA) _(—) _(PIX) line.

In this configuration where V_(AA) _(—) _(PIX)=V_(AA) _(—) _(PIX) _(—) _(CTL)=V_(AA), the pixel reset can be considered to be in between the hard reset and the soft reset modes of operation described above, since the drain voltage and the gate voltage of the reset transistor in a pixel M_(RS) is V_(AA)−V_(th) and V_(AA), respectively, thus M_(RS) operates around a boundary between the linear and saturation regions. If the hard reset mode is required, V_(AA) _(—) _(PIX) _(—) _(CTL) should be slightly lower than V_(AA) _(—) _(PIX), so that the reset transistor inside a pixel M_(RS) operates in the linear region.

Another embodiment is shown in FIG. 16, where both the hard reset and soft reset operation modes can be obtained. A P-MOS transistor 90 (M₂) is added in parallel to n-MOS transistor 80 (M₁). When V_(AA) _(—) _(PIX) _(—) _(CTL) is high (V_(AA)), transistor 90 (M₂) is off, and operation of the circuit is the same as in the previously described embodiment. When V_(AA) _(—) _(PIX) _(—) _(CTL) is low (V_(SS)), transistor 80 (M₁) is off and transistor 90 (M₂) is on. In this case, V_(AA) _(—) _(PIX) _(—) _(COL) becomes V_(AA) _(—) _(pix) and the pixel reset is performed in the soft reset mode, assuming the pulse height of Φ_(RS) is V_(AA).

FIG. 17 shows system 300, a typical processor based system modified to include an image sensor IC 308 including an APS array having compensating circuitry as described above in connection with FIGS. 7-16. Processor based systems exemplify systems of digital circuits that could include an image sensor. Examples of processor based systems include, without limitation, computer systems, camera systems, scanners, machine vision systems, vehicle navigation systems, video telephones, surveillance systems, auto focus systems, star tracker systems, motion detection systems, image stabilization systems, and data compression systems for high-definition television, any of which could utilize the invention.

System 300 includes central processing unit (CPU) 302 that communicates with various devices over bus 304. Some of the devices connected to bus 304 provide communication into and out of system 300, illustratively including input/output (I/O) device 306 and image sensor IC 308. Other devices connected to bus 304 provide memory, illustratively including random access memory (RAM) 310, hard drive 312, and one or more peripheral memory devices such as floppy disk drive 314 and compact disk (CD) drive 316.

Image sensor IC 308 can be implemented as an integrated image sensor circuit on a chip with circuitry to compensate for current gradients, as described above. Image sensor IC 308 may be combined with an image processor, which receives digitized pixel signals from a pixel array and provides digital image output signals. The process can be a CPU, digital signal processor, or microprocessor. The image sensor IC 308 and processor can be combined in a single integrated circuit.

The present invention provides methods and applications for active pixel sensors (APS) having photoreceptors arranged in an array to form a plurality of columns of image sensors. Columns of image sensors have circuitry that compensates for voltage drops in at least one of a voltage supply line and a ground line connecting the plurality of columns of image sensors.

While preferred embodiments of the invention have been described and illustrated above, it should be understood that these are exemplary of the invention and are not to be considered as limiting. Additions, deletions, substitutions, and other modifications can be made without departing from the spirit or scope of the present invention. For example, circuitry can provide no reset capability, or either a hard reset alone, a soft reset alone, or both alone, or in combination with each of the exemplary embodiments for addressing readout circuitry voltage drops. Accordingly, the invention is not to be considered as limited by the foregoing description but is only limited by the scope of the appended claims. 

What is claimed as new and desired to be protected by Letters Patent of the United States is:
 1. An image sensor comprising: a pixel array including a plurality of column lines; a plurality of load transistors, each coupling a respective one of the column lines to a ground line; and a plurality of circuits, each respectively coupled to a gate of a load transistor, the plurality of circuits applying a substantially equal gate-source voltage to the gates of each of the load transistors.
 2. A sensor as in claim 1, wherein the plurality of circuits each comprise a level shift circuit arranged between the ground line and a respective gate of a load transistor.
 3. A sensor as in claim 1, wherein the plurality of circuits each comprise a clamp circuit arranged between the ground line and a respective gate of a load transistor.
 4. An image sensor comprising: a plurality of photoreceptors arranged in an array to form a plurality of columns of image sensors, each column of image sensors having readout circuitry that compensates for voltage drops in a ground line connecting the plurality of columns of image sensors, the readout circuitry comprising a plurality of load transistors each coupling a respective column of image sensors to the ground line, and a plurality of constant gate-source voltage supplying circuits respectively coupled between the ground line and the gates of the load transistors.
 5. A sensor as in claim 4, wherein the plurality of constant gate-source voltage supplying circuits each comprise a level shift circuit connected between the ground line and the gate of a respective load transistor.
 6. A sensor as in claim 4, wherein the plurality of constant gate-source voltage supplying circuits each comprise a clamp circuit.
 7. A sensor as in claim 6, wherein each clamp circuit includes a clamp capacitor charged to a clamping voltage during a period when row select pulses are turned off.
 8. A sensor as in claim 4, wherein each column of image sensors has reset circuitry that compensates for voltage drops in a voltage supply line.
 9. A sensor as in claim 8, wherein the reset circuitry includes a hard reset switch provided in a voltage line to the pixel for hard resetting.
 10. A sensor as in claim 9, wherein the hard reset switch comprises an n-MOS transistor operating in a saturation region.
 11. A sensor as in claim 9, further comprising a soft reset switch provided in parallel to the hard reset switch.
 12. A sensor as in claim 11, wherein the soft reset switch comprises a p-MOS transistor.
 13. A method of operating an image sensor comprising: supplying a constant gate-source voltage to each of the gates of a plurality of load transistors each coupling a respective column of an image sensor array to a ground line such that a bias current flowing through each of the load transistors is substantially the same; selecting a row to obtain a charge signal from a selected row of photoreceptors in the image sensor array; and sampling a signal on each column of the selected row.
 14. The method of claim 13, wherein the act of supplying is performed by a respective level shift circuit coupled between the ground line and a gate a respective load transistor.
 15. A method of operating an image sensor comprising: supplying a constant gate-source voltage to each of the gates of a plurality of load transistors each coupling a respective column of an image sensor array to a ground line such that a bias current flowing through each of the load transistors is substantially the same; selecting a row to obtain a charge signal from a selected row of photoreceptors in the image sensor array; and sampling a signal on each column of the selected row, wherein the act of supplying is performed by a respective level shift circuit coupled between the ground line and a gate a respective load transistor, and wherein an input of the level shift circuit for a given column i is V_(ss) _(—) _(col) _(—) _(i) and an output is V_(ss) _(—) _(col) _(—) _(i)+ΔV, wherein ΔV is a pre-determined voltage associated with the column.
 16. A sensor as in claim 1, wherein the bulk of each load transistor is connected to its source.
 17. A sensor as in claim 2, wherein, for a given column, a respective level shift circuit adds a pre-determined voltage to an input voltage from the ground line.
 18. A sensor as in claim 4, wherein the bulk of each load transistor is connected to its source.
 19. The method of claim 13, wherein the gate-source voltage and a gate-bulk voltage of the load transistor are kept constant even if a voltage on the ground line rises.
 20. A sensor as in claim 1, wherein a voltage supplied by the plurality of circuits maintains a constant bias current through the load transistors.
 21. A sensor as in claim 4, wherein a voltage supplied by the constant gate-source voltage supplying circuits fluctuates to maintain a constant bias current through the load transistors.
 22. The method of claim 15, wherein ΔV for a first column line of the image sensor is different than ΔV for at least one other column line of the image sensor.
 23. An image sensor comprising: a pixel array including a plurality of column lines; a plurality of load transistors, each coupling a respective one of the column lines to a ground line; and having a respective bias current passing therethrough; and the plurality of circuits applying respective gate-source voltages to the gates of the load transistors such that a bias current passing through each of the load transistors is substantially the same. 